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information furnished by analog devices is believed to be accurate and reliable. however, no responsibility is assumed by analog devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. no license is granted by implication or otherwise under any patent or patent rights of analog devices. a ad8302 lf?.7 ghz rf/if gain and phase detector functional block diagram mflt vmag mset pset vphs pflt vref video output ?a inpa ofsa comm ofsb inpb vpos + + 60db log amps (7 detectors) 60db log amps (7 detectors) video output ?b phase detector + bias x3 1.8v ad8302 features measures gain/loss and phase up to 2.7 ghz dual demodulating log amps and phase detector input range ?0 dbm to 0 dbm in a 50 system accurate gain measurement scaling (30 mv/db) typical nonlinearity < 0.5 db accurate phase measurement scaling (10 mv/degree) typical nonlinearity < 1 degree measurement/controller/level comparator modes operates from supply voltages of 2.7 v?.5 v stable 1.8 v reference voltage output small signal envelope bandwidth from dc to 30 mhz applications rf/if pa linearization precise rf power control remote system monitoring and diagnostics return loss/vswr measurements log ratio function for ac signals product description the ad8302 is a fully integrated system for measuring gain/loss and phase in numerous receive, transmit, and instrumentation applications. it requires few external components and a single supply of 2.7 v?.5 v. the ac-coupled i nput signals can range from ?0 dbm to 0 dbm in a 50 ? system, from low frequencies up to 2.7 ghz. the outputs provide an accurate measurement of either gain or loss over a 30 db range scaled to 30 mv/db, and of phase over a 0 ?80 range s caled to 10 mv/degree. both subsystems have an output bandwidth of 30 mhz, which may optionally be reduced by the addition of external f ilter capacitors. the ad8302 can be used in controller mode to force the gain and phase of a signal chain toward predetermined setpoints. the ad8302 comprises a closely matched pair of demodulating logarithmic amplifiers, each having a 60 db measurement range. by taking the difference of their outputs, a measurement of the magnitude ratio or gain betw een the two input signals is available. these signals may even be at different frequencies, allowing the measurement of conversion gain or loss. the ad 8302 may be used to determine absolute signal level by applying the unknown signal to one input and a calibrated ac reference signal to the other. with the output stage feedback connection dis- abled, a com parator may be realized, using the setpoint pins mset and pset to program the thresholds. the signal inputs are single-ended, allowing them to be matched and connected directly to a directional coupler. t heir input impedance is nominally 3 k ? at low frequencies. the ad8302 includes a phase detector of the multiplier type, but with precise phase balance driven by the fully limited signals appearing at the outputs of the two logarithmic amplifiers. thus, the phase accuracy measurement is independent of signal level over a wide range. the phase and gain output voltages are simultaneously available at loadable ground referenced outputs over the standard output range of 0 v to 1.8 v. the output drivers can source or sink up to 8 ma. a loadable, stable reference voltage of 1.8 v is avail- able for precise repositioning of the output range by the user. in controller applications, the connection between the gain output pin vmag and the setpoint control pin mset is broken. the desired setpoint is presented to mset and the vmag control signal drives an appropriate external variable gain device. likewise, the feedback path between the phase output pin vphs and its setpoint control pin pset may be broken to allow operation as a phase controller. the ad8302 is fabricated on analog devices?proprietary, high performance 25 ghz soi complementary bipolar ic process. it is available in a 14-lead tssop package and operates over a ?0 c to +85 c temperature range. an evaluation board is available. one te chn ology way, p.o. box 9106, norwood, ma 02062-9106, u.s.a. t el: 781.329.4700 ?2018 analog devices, inc. all rights reserved. technical support www.analog.com document feedback rev. b
? ad8302?pecifications (t a = 25 c, v s = 5 v, vmag shorted to mset, vphs shorted to pset, 52.3 shunt resistors connected to inpa and inpb, for phase measurement p inpa = p inpb , unless otherwise noted.) parameter conditions min typ max unit overall function input frequency range >0 2700 mhz gain measurement range p in at inpa, p in at inpb = ?0 dbm 30 db phase measurement range in at inpa > in at inpb 90 degree reference voltage output pin vref, ?0 c t a +85 c 1.72 1.8 1.88 v input interface pins inpa and inpb input simplified equivalent circuit to ac ground, f 500 mhz 3 2k ? pf input voltage range ac-coupled (0 dbv = 1 v rms) ?3 ?3 dbv re: 50 ? ?0 0 dbm center of input dynamic range ?3 dbv ?0 dbm magnitude output pin vmag output voltage minimum 20 log (v inpa /v inpb ) = ?0 db 30 mv output voltage maximum 20 log (v inpa /v inpb ) = +30 db 1.8 v center point of output (mcp) v inpa = v inpb 900 mv output current source/sink 8 ma small signal envelope bandwidth pin mflt open 30 mhz slew rate 40 db change, load 20 pf 10 k ? 25 v/ s response time rise time any 20 db change, 10%?0% 50 ns fall time any 20 db change, 90%?0% 60 ns settling time full-scale 60 db change, to 1% settling 300 ns phase output pin vphs output voltage minimum phase difference 180 degrees 30 mv output voltage maximum phase difference 0 degrees 1.8 v phase center point when inpa = inpb 90 900 mv output current drive source/sink 8 ma slew rate 25 v/ s small signal envelope bandwidth 30 mhz response time any 15 degree change, 10%?0% 40 ns 120 degree change c filt = 1 pf, to 1% settling 500 ns 100 mhz magnitude output dynamic range 1 db linearity p ref = ?0 dbm (v ref = ?3 dbv) 58 db 0.5 db linearity p ref = ?0 dbm (v ref = ?3 dbv) 55 db 0.2 db linearity p ref = ?0 dbm (v ref = ?3 dbv) 42 db slope from linear regression 29 mv/db deviation vs. temperature deviation from output at 25 c ?0 c t a +85 c, p inpa = p inpb = ?0 dbm 0.25 db deviation from best fit curve at 25 c ?0 c t a +85 c, p inpa = 25 db, p inpb = ?0 dbm 0.25 db gain measurement balance p inpa = p inpb = ? dbm to ?0 dbm 0.2 db phase output dynamic range less than 1 degree deviation from best fit line 145 degree less than 10% deviation in instantaneous slope 143 degree slope (absolute value) from linear regression about ?0 or +90 10 mv/degree deviation vs. temperature deviation from output at 25 c ?0 c t a +85 c, delta phase = 90 degrees 0.7 degree deviation from best fit curve at 25 c ?0 c t a +85 c, delta phase = 30 degrees 0.7 degree rev. b
? ad8302 parameter conditions min typ max unit 900 mhz magnitude output dynamic range 1 db linearity p ref = ?0 dbm (v ref = ?3 dbv) 58 db 0.5 db linearity p ref = ?0 dbm (v ref = ?3 dbv) 54 db 0.2 db linearity p ref = ?0 dbm (v ref = ?3 dbv) 42 db slope from linear regression 28.7 mv/db deviation vs. temperature deviation from output at 25 c ?0 c t a +85 c, p inpa = p inpb = ?0 dbm 0.25 db deviation from best fit curve at 25 c ?0 c t a +85 c, p inpa = 25 db, p inpb = ?0 dbm 0.25 db gain measurement balance p inpa = p inpb = ? dbm to ?0 dbm 0.2 db phase output dynamic range less than 1 degree deviation from best fit line 143 degree less than 10% deviation in instantaneous slope 143 degree slope (absolute value) from linear regression about ?0 or +90 10.1 mv/degree deviation linear deviation from best fit curve at 25 c ?0 c t a +85 c, delta phase = 90 degrees 0.75 degree ?0 c t a +85 c, delta phase = 30 degrees 0.75 degree phase measurement balance phase @ inpa = phase @ inpb, p in = ? dbm to ?0 dbm 0.8 degree 1900 mhz magnitude output dynamic range 1 db linearity p ref = ?0 dbm (v ref = ?3 dbv) 57 db 0.5 db linearity p ref = ?0 dbm (v ref = ?3 dbv) 54 db 0.2 db linearity p ref = ?0 dbm (v ref = ?3 dbv) 42 db slope from linear regression 27.5 mv/db deviation vs. temperature deviation from output at 25 c ?0 c t a +85 c, p inpa = p inpb = ?0 dbm 0.27 db deviation from best fit curve at 25 c ?0 c t a +85 c, p inpa = 25 db, p inpb = ?0 dbm 0.33 db gain measurement balance p inpa = p inpb = ? dbm to ?0 dbm 0.2 db phase output dynamic range less than 1 degree deviation from best fit line 128 degree less than 10% deviation in instantaneous slope 120 degree slope (absolute value) from linear regression about ?0 or +90 10.2 mv/degree deviation linear deviation from best fit curve at 25 c ?0 c t a +85 c, delta phase = 90 degrees 0.8 degree ?0 c t a +85 c, delta phase = 30 degrees 0.8 degree phase measurement balance phase @ inpa = phase @ inpb, p in = ? dbm to ?0 dbm 1 degree 2200 mhz magnitude output dynamic range 1 db linearity p ref = ?0 dbm (v ref = ?3 dbv) 53 db 0.5 db linearity p ref = ?0 dbm (v ref = ?3 dbv) 51 db 0.2 db linearity p ref = ?0 dbm (v ref = ?3 dbv) 38 db slope from linear regression 27.5 mv/db deviation vs. temperature deviation from output at 25 c ?0 c t a +85 c, p inpa = p inpb = ?0 dbm 0.28 db deviation from best fit curve at 25 c ?0 c t a +85 c, p inpa = 25 db, p inpb = ?0 dbm 0.4 db gain measurement balance p inpa = p inpb = ? dbm to ?0 dbm 0.2 db phase output dynamic range less than 1 degree deviation from best fit line 115 degree less than 10% deviation in instantaneous slope 110 degree slope (absolute value) from linear regression about ?0 or +90 10 mv/degree deviation linear deviation from best fit curve at 25 c ?0 c t a +85 c, delta phase = 90 degrees 0.85 degree ?0 c t a +85 c, delta phase = 30 degrees 0.9 degree reference voltage pin vref output voltage load = 2 k ? 1.7 1.8 1.9 v psrr v s = 2.7 v to 5.5 v 0.25 mv/v output current source/sink (less than 1% change) 5 ma power supply pin vpos supply 2.7 5.0 5.5 v operating current (quiescent) v s = 5 v 19 25 ma ?0 c t a +85 c2127ma specifications subject to change without notice. rev. b
ad8302 ? absolute maximum ratings 1 supply voltage v s . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.5 v pset, mset voltage . . . . . . . . . . . . . . . . . . . . . . v s + 0.3 v inpa, inpb maximum input . . . . . . . . . . . . . . . . . . ? dbv equivalent power re. 50 ? . . . . . . . . . . . . . . . . . . 10 dbm ja 2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 150 c/w maximum junction temperature . . . . . . . . . . . . . . . . 125 c operating temperature range . . . . . . . . . . . ?0 c to +85 c storage temperature range . . . . . . . . . . . . ?5 c to +150 c lead temperature range (soldering 60 sec) . . . . . . . . 300 c notes 1 stresses above those listed under absolute maximum ratings may cause perma- nent damage to the device. this is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. exposure to absolute maximum rating conditions for extended periods may affect device reliability. 2 jedec 1s standard (2-layer) board data. pin configuration top view (not to scale) 1 comm ad8302 inpa ofsa vpos ofsb inpb comm mflt vmag mset vref pset vphs pflt 2 3 4 5 6 7 14 13 12 11 10 9 8 pin function descriptions equivalent pin no. mnemonic function circuit 1, 7 comm device common. connect to low impedance ground. 2 inpa high input impedance to channel a. must be ac-coupled. circuit a 3 ofsa a capacitor to ground at this pin sets the offset compensation filter corner circuit a and provides input decoupling. 4 vpos voltage supply (v s ), 2.7 v to 5.5 v 5 ofsb a capacitor to ground at this pin sets the offset compensation filter corner circuit a and provides input decoupling. 6 inpb input to channel b. same structure as inpa. circuit a 8 pflt low pass filter terminal for the phase output circuit e 9 vphs single-ended output proportional to the phase difference between inpa circuit b and inpb. 10 pset feedback pin for scaling of vphs output voltage in measurement mode. circuit d apply a setpoint voltage for controller mode. 11 vref internally generated reference voltage (1.8 v nominal) circuit c 12 mset feedback pin for scaling of vmag output voltage measurement mode. circuit d accepts a set point voltage in controller mode. 13 vmag single-ended output. output voltage proportional to the decibel ratio of signals applied to inpa and inpb. circuit b 14 mflt low pass filter terminal for the magnitude output circuit e caution esd (electrostatic discharge) sensitive device. electrostatic charges as high as 4000 v readily accumulate on the human body and test equipment and can discharge without detection. although the ad8302 features proprietary esd protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. therefore, proper esd precautions are recommended to avoid performance degradation or loss of functionality. warning! esd sensitive device rev. b
ad8302 ? inpa(inpb) ofsa(ofsb) vpos on to log-amp + comm 10pf 4k 100mv 4k circuit a figure 1. equivalent circuits 2k 750 vpos vmag (vphs) class a-b control 25 comm circuit b vpos 10k 5k vref comm circuit c vpos mset (pset) active loads 10k 10k comm circuit d mflt (pflt) vpos comm 1.5pf circuit e rev. b
ad8302 6 typical performance characteristics magnitude ratio db 2.0 30 vmag v 1.8 1.6 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0 25 20 15 10 5 0 5 1015202530 1900 900 100 2700 2200 tpc 1. magnitude output (vmag) vs. input level ratio (gain) v inpa /v inpb , frequencies 100 mhz, 900 mhz, 1900 mhz, 2200 mhz, 2700 mhz, 25 c, p inpb = 30 dbm, (re: 50 ? ) magnitude ratio db 2.0 30 vmag v 1.8 1.6 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0 25 20 15 10 5 0 5 1015202530 900 100 2200 1900 2700 tpc 2. vmag vs. input level ratio (gain) v inpa /v inpb , frequencies 100 mhz, 900 mhz, 1900 mhz, 2200 mhz, 2700 mhz, p inpa = 30 dbm magnitude ratio db 30 vmag v 1.80 0 20 100 102030 3.0 error in vmag db 2.5 2.0 1.5 1.0 0.5 0.0 0.5 1.0 1.5 3.0 2.0 2.5 1.65 1.50 1.35 1.20 1.05 0.90 0.75 0.60 0.45 0.30 0.15 40c +25c +85c tpc 3. vmag output and log conformance vs. input level ratio (gain), frequency 100 mhz, 40 c, +25 c, and +85 c, reference level = 30 dbm magnitude ratio db 30 vmag v 1.80 0 20 10 0 10 20 30 3.0 error in vmag db 2.5 2.0 1.5 1.0 0.5 0.0 0.5 1.0 1.5 3.0 2.0 2.5 1.65 1.50 1.35 1.20 1.05 0.90 0.75 0.60 0.45 0.30 0.15 +25c +85c 40c tpc 4. vmag and log conformance vs. input level ratio (gain), frequency 900 mhz, 40 c, +25 c, and +85 c, reference level = 30 dbm magnitude ratio db 30 vmag v 1.80 1.65 0 20 10 0 10 20 30 3.0 error in vmag db 2.5 2.0 1.5 1.0 0.5 0.0 0.5 1.0 1.5 3.0 2.0 2.5 1.50 1.35 1.20 1.05 0.90 0.75 0.60 0.45 0.30 0.15 +85c 40c +25c tpc 5. vmag and log conformance vs. input level ratio (gain), frequency 1900 mhz, 40 c, +25 c, and +85 c, reference level = 30 dbm magnitude ratio db 30 vmag v 20 10 0 10 20 30 3.0 error in vmag db 2.5 2.0 1.5 1.0 0.5 0.0 0.5 1.0 1.5 3.0 2.0 2.5 1.80 1.65 0 1.50 1.35 1.20 1.05 0.90 0.75 0.60 0.45 0.30 0.15 +25c +85c 40c tpc 6. vmag output and log conformance vs. input level r atio (gain), frequency 2200 mhz, 40 c, +25 c, and +85 c, reference level = 30 dbm (v s = 5 v, v inpb is the reference input and v inpa is swept, unless otherwise noted. all references to dbm are referred to 50 . for the phase output curves, the input signal levels are equal, unless otherwise noted.) rev. b
ad8302 7 magnitude ratio db 30 error in vmag db 1.5 1.0 0.5 0.0 0.5 1.0 1.5 2.0 2.5 3.0 25 20 15 10 5 0 5 1015202530 2.0 3.0 2.5 40 c +85 c +25 c 40 c +85 c tpc 7. distribution of magnitude error vs. input level ratio (gain), three sigma to either side of mean, frequency 900 mhz, 40 c, +25 c, and +85 c, refer- ence level = 30 dbm magnitude ratio db 30 error in vmag db 1.5 1.0 0.5 0.0 0.5 1.0 1.5 2.0 2.5 3.0 25 20 15 10 5 0 5 1015202530 2.0 3.0 2.5 40 c +85 c +25 c +85 c 40 c tpc 8. distribution of error vs. input level ratio (gain), three sigma to either side of mean, frequency 1900 mhz, 40 c, +25 c, and +85 c, reference level = 30 dbm magnitude ratio db 30 error in vmag db 1.5 1.0 0.5 0.0 0.5 1.0 1.5 2.0 2.5 3.0 25 20 15 10 5 0 5 1015202530 2.0 3.0 2.5 40 c +85 c +25 c +85 c 40 c tpc 9. distribution of magnitude error vs. input level ratio (gain), three sigma to either side of mean, frequency 2200 mhz, temperatures 40 c, +25 c, and +85 c, reference level = 30 dbm magnitude ratio db 30 vmag v 1.8 1.6 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0.0 25 20 15 10 5 0 5 1015202530 2.0 tpc 10. distribution of vmag vs. input level ratio (gain), three sigma to either side of mean, frequency 1900 mhz, temperatures between 40 c and +85 c, reference level = 30 dbm magnitude ratio db 30 vmag v 1.2 1.0 0.8 0.6 0.4 0.2 0.0 20 100 102030 1.4 1.8 1.6 45dbm error in vmag db 1.5 1.0 0.5 0.0 0.5 1.0 1.5 2.0 3.0 2.5 2.0 2.5 3.0 30dbm 45dbm 30dbm 15dbm 15dbm tpc 11. vmag output and log conformance vs. input level ratio (gain), reference level = 15 dbm, 30 dbm, and 45 dbm, frequency 1900 mhz input level dbm 65 vmag v 0.90 0.85 0.80 0.75 60 55 50 45 40 35 30 0.95 1.05 1.00 25 20 15 10 50 p inpa = p inpb p inpa = p inpb 5db p inpa = p inpb + 5db 1.10 tpc 12. vmag output vs. input level for p inpa = p inpb , p inpa = p inpb + 5 db, p inpa = p inpb 5 db, frequency 1900 mhz rev. b
ad8302 8 frequency mhz vmag v 200 400 600 800 1000 1200 1400 1.06 1600 1800 2000 2200 0 1.04 1.02 1.00 0.98 0.96 0.94 0.92 0.90 0.88 0.86 0.84 0.82 0.80 0.78 0.76 0.74 p inpa = p inpb + 5db p inpa = p inpb p inpa = p inpb 5db tpc 13. vmag output vs. frequency, for p inpa = p inpb , p inpa = p inpb + 5 db, and p inpa = p inpb 5 db, p inpb = 30 dbm temperature c change in slope mv 40 200 20406080 0.4 85 0.2 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 tpc 14. change in vmag slope vs. temperature, three sigma to either side of mean, frequencies 1900 mhz temperature c vmag mv 5 40 30 20 10 0 10 20 5 25 15 30 40 50 60 20 10 0 10 15 20 25 70 80 90 tpc 15. change in center point of magnitude output (mcp) vs. temperature, three sigma to either side of mean, frequencies 1900 mhz 0.80 0.85 0.90 12 18 15 0.95 9 6 3 0 1.00 percent mcp v tpc 16. center point of magnitude output (mcp) distribution frequencies 900 mhz, 17,000 units 27.0 27.5 28.0 28.5 12 18 15 29.0 9 6 3 0 29.5 30.0 percent vmag slope mv/db tpc 17. vmag slope, frequency 900 mhz, 17,000 units frequency mhz slope of vmag v 0 0.032 0.030 0.028 0.026 0.024 200 400 600 800 1000 1200 1400 1600 1800 2000 2200 2400 2600 2800 tpc 18. vmag slope vs. frequency rev. b
ad8302 9 25ns horizontal 20mv per vertical division tpc 19. magnitude output response to 4 db step, for p inpb = 30 dbm, p inpa = 32 dbm to 28 dbm, frequency 1900 mhz, no filter capacitor 1.00s horizontal 20mv per vertical division tpc 20. magnitude output response to 4 db step, for p inpb = 30 dbm, p inpa = 32 dbm to 28 dbm, frequency 1900 mhz, 1 nf filter capacitor 100ns horizontal 200mv per vertical division tpc 21. magnitude output response to 40 db step, for p inpb = 30 dbm, p inpa = 50 dbm to 10 dbm, supply 5 v, frequency 1900 mhz, no filter capacitor frequency hz vmag nv/ hz 1k 10k 10000 100k 1m 10m 100m 1000 100 10 input 50dbm input 30dbm input 10dbm tpc 22. magnitude output noise spectral density, p inpa = p inpb = 10 dbm, 30 dbm, 50 dbm, no filter capacitor frequency hz vmag nv/ hz 1k 10k 10000 100k 1m 10m 100m 1000 100 10 input 50dbm input 30dbm input 10dbm tpc 23. magnitude output noise spectral density, p inpa = p inpb = 10 dbm, 30 dbm, 50 dbm, with filter capacitor, c = 1 nf magnitude ratio db vmag (peak-to-peak) v 25 20 0.18 15 10 25 5 0 5 101520 100 0.16 0.14 0.12 0.10 0.08 0.06 0.04 0.02 0.00 900 1900 2200 2700 tpc 24. vmag peak-to-peak output induced by sweeping phase difference through 360 degrees vs. magnitude ratio, frequencies 100 mhz, 900 mhz, 1900 mhz, 2200 mhz, and 2700 mhz rev. b
ad8302 10 phase difference degrees phase out v 180 140 1.8 100 60 20 20 60 100 140 180 1.6 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0.0 900mhz 100mhz 1900mhz 2200mhz 2700mhz tpc 25. phase output (vphs) vs. input phase difference, input levels 30 dbm, frequencies 100 mhz, 900 mhz, 1900 mhz, 2200 mhz, supply 5 v, 2700 mhz phase difference degrees phase out v 180 150 1.80 120 90 60 300 306090 1.62 1.44 1.26 1.08 0.90 0.72 0.54 0.36 0.00 120 150 180 0.18 error degrees 10 8 6 4 2 0 2 4 6 10 8 tpc 26. vphs output and nonlinearity vs. input phase difference, input levels 30 dbm, frequency 100 mhz phase difference degrees phase out v 180 150 1.80 120 90 60 300 306090 1.62 1.44 1.26 1.08 0.90 0.72 0.54 0.36 0.00 120 150 180 0.18 error degrees 10 8 6 4 2 0 2 4 6 10 8 tpc 27. vphs output and nonlinearity vs. input phase difference, input levels 30 dbm, frequency 900 mhz phase difference degrees phase out v 180 150 1.80 120 90 60 300 306090 1.62 1.44 1.26 1.08 0.90 0.72 0.54 0.36 0.00 120 150 180 0.18 error degrees 10 8 6 4 2 0 2 4 6 10 8 tpc 28. vphs output and nonlinearity vs. input phase difference, input levels 30 dbm, frequency 1900 mhz phase difference degrees phase out v 180 150 1.80 120 90 60 300 306090 1.62 1.44 1.26 1.08 0.90 0.72 0.54 0.36 0.00 120 150 180 0.18 error degrees 10 8 6 4 2 0 2 4 6 10 8 tpc 29. vphs output and nonlinearity vs. input phase difference, input levels 30 dbm, frequency 2200 mhz phase difference degrees error degrees 180 150 10 120 90 60 300 306090 8 6 4 2 0 2 4 6 10 120 150 180 8 40c +85c +25c tpc 30. distribution of vphs error vs. input phase differ- ence, three sigma to either side of mean, frequency 900 mhz, 40 c, +25 c, and +85 c, input levels 30 dbm rev. b
ad8302 11 phase difference degrees error degrees 180 150 10 120 90 60 300 306090 8 6 4 2 0 2 4 6 10 120 150 180 8 40c +85c +25c tpc 31. distribution of vphs error vs. input phase difference, three sigma to either side of mean, frequency 1900 mhz, 40 c, +25 c, and +85 c, supply 5 v, input levels p inpa = p inpb = 30 dbm phase difference degrees error degrees 180 150 10 120 90 60 300 306090 8 6 4 2 0 2 4 6 10 120 150 180 8 40c +85c +25c tpc 32. distribution of vphs error vs. input phase differ- ence, three sigma to either side of mean, frequency 2200 mhz, 40 c, +25 c, and +85 c, input levels 30 dbm phase difference degrees vphs v 180 150 120 90 60 300 306090 1.8 1.6 1.4 1.2 1.0 0.8 0.6 0.4 0.0 120 150 180 0.2 tpc 33. distribution of vphs vs. input phase differ- ence, three sigma to either side of mean, frequency 900 mhz, temperature between 40 c and +85 c, input levels 30 dbm temperature c change in vphs slope mv 40 30 20 100 1020304050 0.35 60 80 90 70 0.30 0.25 0.20 0.15 0.10 0.05 0.00 0.05 0.10 0.15 mean +3 sigma mean 3 sigma tpc 34. change in vphs slope vs. temperature, three sigma to either side of mean, frequency 1900 mhz vphs mv/degree percent 40 30 20 100 1020304050 40 60 80 90 +3 sigma 3 sigma 70 35 30 25 20 15 10 5 0 5 10 tpc 35. change in phase center point (pcp) vs. temperature, three sigma to either side of mean, frequency 1900 mhz pcp v percent 0.75 0.80 0.85 0.90 0.95 18 15 12 9 6 0 1.00 1.05 3 tpc 36. phase center point (pcp) distribution, frequency 900 mhz, 17,000 units rev. b
ad8302 12 vphs mv/degree percent 9.5 9.7 9.9 10.1 10.3 10.5 10.7 10.9 0 2 4 6 8 10 12 14 11.1 16 tpc 37. vphs slope distribution, frequency 900 mhz 50ns horizontal 10mv per vertical division tpc 38. vphs output response to 4 step with nominal phase shift of 90 , input levels 30 dbm, frequency 1900 mhz, 25 c, 1 pf filter capacitor 2 s horizontal 10mv per vertical division tpc 39. vphs output response to 4 step with nominal phase shift of 90 , input levels p inpa = p inpb = 30 dbm, supply 5 v, frequency 1900 mhz, 25 c, with 100 pf filter capacitor 50ns horizontal 100mv per vertical division tpc 40. vphs output response to 40 step with nominal phase shift of 90 , input levels p inpa = p inpb = 30 dbm, frequency 1900 mhz,1 pf filter capacitor frequency hz vphs nv/ hz 1k 10000 1000 100 10 10k 100k 1m 10m 100m input 50dbm input 30dbm input 10dbm tpc 41. vphs output noise spectral density vs. frequency, p inpa = 30 dbm, p inpb = 10 dbm, 30 dbm, 50 dbm, and 90 input phase difference phase difference degrees phase out v 180 150 1.80 120 90 60 300 306090 1.62 1.44 1.26 1.08 0.90 0.72 0.54 0.36 0.00 120 150 180 0.18 p inpa = 45dbm p inpa = 15dbm p inpa = 30dbm tpc 42. phase output vs. input phase difference, p inpa = p inpb , p inpa = p inpb + 15 db, p inpa = p inpb 15 db, frequency 900 mhz rev. b
ad8302 13 phase difference degrees absolute value of vphs instantaneous slope mv 180 150 12 120 90 60 300 306090 10 8 6 4 2 0 120 150 180 p inpa = 30dbm p inpa = 45dbm p inpa = 15dbm tpc 43. phase output instantaneous slope, p inpa = p inpb , p inpa = p inpb + 15 db, p inpa = p inpb 15 db, frequency 900 mhz phase difference degrees phase out v 180 150 1.80 120 90 60 300 306090 1.62 1.44 1.26 1.08 0.90 0.72 0.54 0.36 0.00 120 150 180 0.18 p inpa = 40dbm p inpa = 20dbm p inpa = 30dbm tpc 44. phase output vs. input phase difference, p inpa = p inpb , p inpa = p inpb + 10 db, p inpa = p inpb 10 db, frequency 1900 mhz, supply 5 v phase difference degrees absolute value of vphs instantaneous slope mv 180 150 12 120 90 60 300 306090 10 8 6 4 0 120 150 180 2 p inpa = 40dbm p inpa = 20dbm p inpa = 30dbm tpc 45. phase output instantaneous slope, p inpa = p inpb , p inpa = p inpb + 10 db, p inpa = p inpb 10 db, frequency 1900 mhz, supply 5 v phase difference degrees phase out v 180 150 1.80 120 90 60 300 306090 1.62 120 150 180 p inpa = 40dbm p inpa = 20dbm p inpa = 30dbm 1.44 1.26 1.08 0.90 0.72 0.54 0.36 0.18 0.00 tpc 46. phase output vs. input phase difference, p inpa = p inpb , p inpa = p inpb + 10 db, p inpa = p inpb 10 db, frequency 2200 mhz phase difference degrees 180 150 12 120 90 60 30 0 30 60 90 120 150 180 p inpa = 40dbm p inpa = 20dbm p inpa = 30dbm 10 8 6 4 2 0 absolute value of vphs instantaneous slope mv tpc 47. phase output instantaneous slope, p inpa = p inpb , p inpa = p inpb + 10 db, p inpa = p inpb 10 db, frequency 2200 mhz real shunt z () frequency mhz resistance 0 4000 500 1000 1500 2000 2500 3500 0 3000 2500 2000 1500 1000 500 capacitance pf 4.0 3.5 0.0 3.0 2.5 2.0 1.5 1.0 0.5 shunt c shunt r capacitance shunt z (pf) tpc 48. input impedance, modeled as shunt r in parallel with shunt c rev. b
ad8302 14 temperature c vref mv 40 30 8 20 100 10203040506070 90 4 2 0 2 4 6 6 80 tpc 49. change in vref vs. temperature, three sigma to either side of mean frequency hz 1k 120 80 60 40 20 0 100 10k 100k 1m 10m 100m noise nv/ hz tpc 50. vref output noise spectral density vs. frequency vref v 1.74 18 12 9 6 3 0 15 1.78 1.82 1.84 1.86 1.88 percent 1.76 1.80 tpc 51. vref distribution, 17,000 units rev. b
ad8302 15 general description and theory the ad8302 measures the magnitude ratio, defined here as gain, and phase difference between two signals. a pair of matched logarithmic amplifiers provide the measurement, and their hard-limited outputs drive the phase detector. basic theory logarithmic amplifiers (log amps) provide a logarithmic com- pression function that converts a large range of input signal levels to a compact decibel-scaled output. the general math- ematical form is: vv vv out slp in z = () log / (1) where v in is the input voltage, v z is called the intercept (voltage), and v slp is called the slope (voltage). it is assumed throughout that log(x) represents the log10(x) function. v slp is thus the volts/decade, and since a decade of voltage corresponds to 20 db, v slp /20 is the volts/db. v z is the value of input signal that results in an output of zero and need not correspond to a physically realizable part of the log amp signal range. while the slope is fundamentally a characteristic of the log amp, the intercept is a function of the input waveform as well. 1 furthermore, the intercept is typica lly more sensitive to tem- perature and frequency than the slope. when single log amps are used for power measurement, this variability introduces errors into the absolute accuracy of the measurement since the intercept represents a reference level. the ad8302 takes the difference in the output of two identical log amps, each driven by signals of similar waveforms but at different levels. since subtraction in the logarithmic domain corresponds to a ratio in the linear domain, the resulting output becomes: vv vv mag slp ina inb = () log / (2) where v ina and v inb are the input voltages, v mag is the output corresponding to the magnitude of the signal level difference, and v slp is the slope. note that the intercept, v z , has dropped out. unlike the measurement of power, when measuring a dimen- sion less quantity such as relative signal level, no independent reference or intercept need be invoked. in essence, one signal serves as the intercept for the other. variations in intercept due to frequency, process, temperature, and supply voltage affect both channels identically and hence do not affect the difference. this technique depends on the two log amps being well matched in slope and intercept to ensure cancellation. this is the case for an integrated pair of log amps. n ote that if the two signals have different waveforms (e.g., diffe rent peak-to-average ratios) or different frequencies, an interc ept difference may appear, intro- ducing a systematic offset. the log amp structure consists of a cascade of linear/limiting gain stages with demodulating detectors. further details about the structure and function of log amps can be found in data sheets for other log amps produced by analog devices. 2 the output of the final stage of a log amp is a fully limited signal over most of the input dynamic range. the limited outputs from both log amps drive an exclusive-or style digital phase detector. operating strictly on the relative zero-crossings of the limited sig- nals, the extracted phase difference is independent of the original input signal levels. the phase output has the general form: vvv v phs ina inb = () ? () [] ? (3) where v is the phase slope in mv/degree and is each signal s relative phase in degrees. structure the general form of the ad8302 is shown in figure 2. the major blocks consist of two demodulating log amps, a phase detector, output amplifiers, a biasing cell, and an output refer- ence voltage buffer. the log amps and phase detector process the high frequency signals and deliver the gain and phase infor- mation in current form to the output amplifiers. the output amplifiers determine the final gain and phase scaling. external filter capacitors set the averaging time constants for the respec- tive outputs. the reference buffer provides a 1.80 v reference voltage that tracks the internal scaling constants. mflt vmag mset pset vphs pflt vref video output e a inpa ofsa comm ofsb inpb vpos + e + e 60db log amps (7 detectors) 60db log amps (7 detectors) video output e b phase detector + e bias x3 1.8v figure 2. general structure each log amp consists of a cascade of six 10 db gain stages with seven associated detectors. the individual gain stages have 3 db bandwidths in excess of 5 ghz. the signal path is fully differen- tial to minimize the effect of common-mode signals and noise. since there is a total of 60 db of cascaded gain, slight dc offsets can cause limiting of the latter stages, which may cause mea- surement errors for small signals. this is corrected by a feedback loop. the nominal high-pass corner frequency, f hp , of this loop is set internally at 200 mhz but can be lowered by adding external capacitance to the ofsa and ofsb pins. signals at frequencies well below the high-pass corner are indistinguishable from dc offsets and are also nulled. the difference in the log amp out- puts is performed in the current domain, yielding by analogy to equation 2: ii vv la slp ina inb = () log / (4) where i la and i slp are the output current difference and the characteristic slope (current) of the log amps, respectively. the slope is derived from an accurate reference designed to be insen- sitive to temperature and supply voltage. the phase detector uses a fully symmetric structure with respect to its two inputs to maintain balanced delays along both signal paths. fully differential signaling again minimizes the sensitivity to common-mode perturbations. the current-mode equivalent to equation 3 is: iiv v pd ina inb = () ? () ? [] ? 90 (5) where i pd and i are the output current and characteristic slope associated with the phase detector, respectively. the slope is derived from the same reference as the log amp slope. notes 1 see the data sheet for the ad640 for a description of the effect of waveform on the intercept of log amps. 2 for example, see the data sheet for the ad8307. rev. b
ad8302 16 note that by convention, the phase difference is taken in the range from 180 to +180 . since this style of phase detector does not distinguish between 90 , it is considered to have an unambiguous 180 phase difference range that can be either 0 to +180 centered at +90 or 0 to 180 centered at 90 . the basic structure of both output interfaces is shown in figure 3. it accepts a setpoint input and includes an internal integrating/averag- ing capacitor and a buffer amplifier with gain k. external access to these setpoints provides for several modes of operation and enables flexible tailoring of the gain and phase transfer characteristics. the setpoint interface block, characterized by a transresistance r f , gener- ates a current proportional to the voltage presented to its input pin, mset or pset. a precise offset voltage of 900 mv is introduced internally to establish the center-point (v cp ) for the gain and phase functions, i.e., the setpoint voltage that corresponds to a gain of 0 db and a phase difference of 90 . this setpoint current is subtracted from the signal current, i in , coming from the log amps in the gain channel or from the phase detector in the phase channel. the result- ing difference is integrated on the averaging capacitors at either pin mflt or pflt and then buffered by the output amplifier to the respective output pins, vmag and vphs. with this open-loop arrangement, the output voltage is a simple integration of the differ- ence between the measured gain/phase and the desired setpoint: vriist out f in fb =? ()() / (6) where i fb is the feedback current equal to ( v set ?v cp )/ r f , v set is the setpoint input, and t is the integration time constant equal to r f c ave /k, where c ave is the parallel combination of the inter- nal 1.5 pf and the external capacitor c flt . k r f mset/pset 20k + + v cp = 900mv 1.5pf c flt mflt/pflt vmag/vphs i in = i la or i pd i fb + e figure 3. simplified block diagram of the output interface basic connections measurement mode the basic function of the ad8302 is the direct measurement of gain and phase. when the output pins, vmag and vphs, are connected directly to the feedback setpoint input pins, mset and pset, the default slopes and center points are invoked. this basic connection shown in figure 4 is termed the measurement mode. the current from the setpoint interface is forced by the integrator to be equal to the signal currents coming from the log amps and phase detector. the closed loop transfer function is thus given by: virv/st out in f cp =+ () + () 1 (7) the time constant t represents the single-pole response to the enve- lope of the db-scaled gain and the degree-scaled phase functions. a small internal capacitor sets the maximum envelope bandwidth to approximately 30 mhz. if no external c flt is used, the ad8302 can follow the gain and phase envelopes within this bandwidth. if longer averaging is desired, c flt can be added as necessary accord- ing to t (ns) = 3.3 c ave (pf). for best transient response with minimal overshoot, it is recommended that 1 pf minimum value external capacitors be added to the mflt and pflt pins. 1 comm mflt 14 inpa vmag 21 3 ofsa mset 31 2 vpos vref 41 1 ofsb pset 51 0 inpb vphs 69 comm pflt 78 ad8302 c2 v mag v phs c8 c1 c4 c6 c5 r1 r2 v ina v inb vp c7 r4 c3 figure 4. basic connections in measurement mode with 30 mv/db and 10 mv/degree scaling in the low frequency limit, the gain and phase transfer functions given in equations 4 and 5 become: vrivvvor mag f slp ina inb cp = () + log / (8a) vri ppv mag f slp ina inb cp = () ? () + /20 (8b) vriv v v phs f ina inb cp = () ? () () + || ? 90 (9) which are illustrated in figure 5. in equation 8b, p ina and p inb are the power in dbm equivalent to v ina and v inb at a specified refer- ence impedance. for the gain function, the slope represented by r f i slp is 600 mv/decade or, dividing by 20 db/decade, 30 mv/db. with a center point of 900 mv for 0 db gain, a range of 30 db to +30 db covers the full-scale swing from 0 v to 1.8 v. for the phase function, the slope represented by r f i is 10 mv/degree. with a center point of 900 mv for 90 , a range of 0 to 180 covers the full- scale swing from 1.8 v to 0 v. the range of 0 to 180 covers the same full-scale swing but with the opposite slope. 1.8v 900mv 0v v mag v phs 30mv/db v cp magnitude ratio e db e30 0 +30 1.8v 900mv 0v phase difference e degrees +10mv/deg e10mv/deg v cp e180 e90 0 90 180 figure 5. idealized transfer characteristics for the gain and phase measurement mode rev. b
ad8302 17 interfacing to the input channels the single-ended input interfaces for both channels are identical. each consists of a driving pin, inpa and inpb, and an ac- grounding pin, ofsa and ofsb. all four pins are internally dc-biased at about 100 mv from the positive supply and should be externally ac-coupled to the input signals and to ground. for the signal pins, the coupling capacitor should offer negligible impedance at the signal frequency. for the grounding pins, the coupling capacitor has two functions: it provides ac grounding and sets the high-pass corner frequency for the internal offset compensation loop. there is an internal 10 pf capacitor to ground that sets the maximum corner to approximately 200 mhz. the corner can be lowered according the formula f hp (mhz) = 2/c c (nf), where c c is the total capacitance from ofsa or ofsb to ground, including the internal 10 pf. the input impedance to inpa and inpb is a function of frequency, the offset compensation capacitor, and package parasitics. at moderate frequencies above f hp , the input network can be approximated by a shunt 3 k ? resistor in parallel with a 2 pf capacitor. at higher frequencies, the shunt resistance decreases to approximately 500 ? . the smith chart in figure 6 shows the input impedance over the frequency range 100 mhz to 3 ghz. 2.2ghz 2.7ghz 3.0ghz 1.8ghz 900mhz 100mhz figure 6. smith chart showing the input impedance of a single channel from 100 mhz to 3 ghz a broadband resistive termination on the signal side of the coupling capacitors can be used to match to a given source i mpedance. the value of the termination resistor, r t , is determined by: rrrr r t in s in s =? () / (10) where r in is the input resistance and r s the source impedance. at higher frequencies, a reactive, narrow-band match might be desirable to tune out the reactive portion of the input impedance. an important attribute of the two-log-amp architecture is t hat if both channels are at the same frequency and have the same input network, then impedance mismatches and reflection losses become essentially common-mode and hence do not impact the relative gain and phase measurement. however, mismatches in these external components can result in measurement errors. dynamic range the maximum measurement range for the gain subsystem is lim- ited to a total of 60 db distributed from 30 db to +30 db. this means that both gain and attenuation can be measured. the limits are determined by the minimum and maximum levels that each individual log amp can detect. in the ad8302, each log amp can detect inputs ranging from 73 dbv [(223 v, 60 dbm re: 50 ? to 13 dbv (223 mv, 0 dbm re: 50 ? )]. note that log amps respond to voltages and not power. an equivalent power can be inferred given an impedance level, e.g., to convert from dbv to dbm in a 50 ? system, simply add 13 db. to cover the entire range, it is necessary to apply a reference level to one log amp that corresponds precisely to its midrange. in the ad8302, this level is at 43 dbv, which corresponds to 30 dbm in a 50 ? environment. the other channel can now sweep from its low end, 30 db below midrange, to its high end, 30 db above midrange. if the reference is displaced from midrange, some measurement range will be lost at the extremes. this can occur either if the log amps run out of range or if the rails at ground or 1.8 v are reached. figure 7 illustrates the effect of the reference channel level place ment. if the reference is chosen lower than midrange by 10 db, then the lower limit will be at 20 db rather than 30 db. if the reference chosen is higher by 10 db, the upper limit will be 20 db rather than 30 db. gain measurement range e db e30 0 +30 1.80 0.90 vmag e v max range for v ref = v ref opt v ref > v ref opt v ref < v ref opt figure 7. the effect of offsetting the reference level is to reduce the maximum dynamic range the phase measurement range is of 0 to 180 . for phase differ- ences of 0 to 180 , the transfer characteristics are mirrored as shown in figure 5, with a slope of the opposite sign. the phase detector responds to the relative position of the zero crossings between the two input channels. at higher frequencies, the finite rise and fall times of the amplitude limited inputs create an ambiguous situation that leads to inaccessible dead zones at the 0 and 180 limits. for maximum phase difference coverage, the reference phase difference should be set to 90 . rev. b
ad8302 18 cross modulation of magnitude and phase at high frequencies, unintentional cross coupling between signals in channels a and b inevitably occurs due to on-chip and board- level parasitics. when the two signals presented to the ad8302 inputs are at very different levels, the cross coupling introduces cross modulation of the phase and magnitude responses. if the two signals are held at the same relative levels and the phase between them is modulated then only the phase output should respond. due to phase-to-amplitude cross modulation, the magnitude out- put shows a residual response. a similar effect occurs when the relative phase is held constant while the magnitude difference is modulated, i.e., an expected magnitude response and a residual phase response are observed due to amplitude-to-phase cross modulation. the point where these effects are noticeable depends on the signal frequency and the magnitude of the difference. typi- cally, for differences <20 db, the effects of cross modulation are negligible at 900 mhz. modifying the slope and center point the default slope and center point values can be modified with the addition of external resistors. since the output interface blocks are generalized for both magnitude and phase functions, the scaling modification techniques are equally valid for both outputs. figure 8 demonstrates how a simple voltage divider from the vmag and vphs pins to the mset and pset pins can be used to modify the slope. the increase in slope is given by 1 + r1/(r2 20 k ? ). note that it may be necessary to account for the mset and pset input impedance of 20 k ? which has a 20% manufacturing tolerance. as is generally true in such feedback systems, envelope bandwidth is decreased and the output noise transferred from the input is increased by the same factor. for example, by selecting r1 and r2 to be 10 k ? and 20 k ? , respectively, gain slope increases from the nominal 30 mv/db by a factor of 2 to 60 mv/db. the range is reduced by a factor of 2 and the new center point is at 15 db, i.e., the range now extends from 30 db, corresponding to v mag = 0 v, to 0 db, corresponding to v mag = 1.8 v. new slope = 30mv/db 1 r1 r2 || r20k vmag mset 20k r1 r2 figure 8. increasing the slope requires the inclusion of a voltage divider repositioning the center point back to its original value of 0 db simply requires that an appropriate voltage be applied to the grounded side of the lower resistor in the voltage divider. this voltage may be provided externally or derived from the internal reference voltage on pin vref. for the specific choice of r2 = 20 k ? , the center point is easily readjusted to 0 db by connecting the vref pin directly to the lower pin of r2 as shown in figure 9. the increase in slope is now simplified to 1 + r1/10 k ? . since this 1.80 v reference voltage is derived from the same band gap reference that determines the nominal center point, their tracking with temperature, supply, and part-to-part variations should be better in comparison to a fixed external voltage. if the center point is shifted to 0 db in the previous example w here the slope was doubled, then the range spans from 15 db at v mag = 0 v to 15 db at v mag = 1.8 v. 1 r1 10k new slope = 30mv/db vmag mset 20k r1 20k vref figure 9. the center point is repositioned with the help of the internal reference voltage of 1.80 v comparator and controller modes the ad8302 can also operate in a comparator mode if used in the arrangement shown in figure 10 where the dut is the element to be evaluated. the vmag and vphs pins are no longer connected to mset and pset. the trip-point thresholds for the gain and phase difference comparison are determined by the voltages applied to pins mset and pset according to: v v mv db gain db mv mset sp () ( ) = + 30 900 (11) v v mv phase mv pset sp () | ()| =? ( ) + 10 90 900 (12) where gain sp ( db ) and phase sp ( ) are the desired gain and phase thresholds. if the actual gain and phase between the two input channels differ from these thresholds, the v mag and v phs outputs toggle like comparators, i.e., 18 0 . v if gain gain v v if gain gain sp mag sp > = < (13) 18 0 . v if phase phase v v if phase phase sp phs sp > = < (14) v mag v mset v pset v phs 1 comm mflt 14 inpa vmag 21 3 ofsa mset 31 2 vpos vref 41 1 ofsb pset 51 0 inpb vphs 69 comm pflt 78 ad8302 c2 c8 c1 c4 c6 c5 r1 r2 v ina v inb vp c7 r4 c3 figure 10. disconnecting the feedback to the setpoint controls, the ad8302 operates in comparator mode / / rev. b
ad8302 19 the comparator mode can be turned into a controller mode by closing the loop around the vmag and vphs outputs. figure 11 i llustrates a closed loop controller that stabilizes the gain and phase of a dut with gain and phase adjustment elements. if vmag and vphs are properly conditioned to drive gain and phase adjustment blocks preceding the dut, the actual gain and phase of the dut will be forced toward the prescribed setpoint gain and phase given in equations 11 and 12. these are essentially agc and apc loops. note that as with all control loops of this kind, loop dynamics and appropriate interfaces all must be considered in more detail. mag setpoint phase setpoint vmag mset pset vphs inpa inpb mag ad8302 figure 11. by applying overall feedback to a dut via external gain and phase adjusters, the ad8302 acts as a controller applications measuring amplifier gain and compression the most fundamental application of ad8302 is the monitoring of the gain and phase response of a functional circuit block such as an amplifier or a mixer. as illustrated in figure 12, directional couplers, dc b and dc a , sample the input and output signals of the black box dut. the attenuators ensure that the signal levels presented to the ad8302 fall within its dynamic range. from the discussion in the dynamic range section, the optimal choice places both channels at p opt = 30 dbm referenced to 50 ? , which corresponds to 43 dbv. to achieve this, the combination of coupling factor and attenuation are given by: clp p b b in opt += ? (15) (16) where c b and c a are the coupling coefficients, l b and l a are the attenuation factors, and gain nom is the nominal dut gain. if identical couplers are used for both ports, then the difference in the two attenuators compensates for the nominal dut gain. when the actual gain is nominal, the vmag output is 900 mv, corresponding to 0 db. variations from nominal gain appear as a deviation from 900 mv or 0 db with a 30 mv/db scaling. depending on the nominal insertion phase associated with dut, the phase measurement may require a fixed phase shift in series with one of the channels to bring the nominal phase difference presented to the ad8302 near the optimal 90 point. when the insertion phase is nominal, the vphs output is 900 mv. deviat ions from the nominal are reported with a 10 mv/degree scaling. table i gives suggested component values for the measurement of an amplifier with a nominal gain of 10 db and an input power of 10 dbm. atten a dc a atten b dc b r5 h r6 h black box output input 1 comm mflt 14 inpa vmag 21 3 ofsa mset 31 2 vpos vref 41 1 ofsb pset 51 0 inpb vphs 69 comm pflt 78 ad8302 c2 c8 c1 c4 c6 c5 r1 r2 vp c7 r4 c3 figure 12. using the ad8302 to measure the gain and insertion phase of an amplifier or mixer table i. component values for measuring a 10 db amplifier with an input power of ?0 dbm component value quantity r1, r2 52.3 ? 2 r5, r6 100 ? 2 c1, c4, c5, c6 0.001 f4 c2, c8 open c3 100 pf 1 c7 0.1 f1 attena 10 db (see text) 1 attenb 1 db (see text) 1 dc a , dc b 20 db 2 the gain measurement application can also monitor gain and phase distortion in the form of am-am (gain compression) and am-pm conversion. in this case, the nominal gain and phase corresponds to those at low input signal levels. as the input level is increased, output compression and excess phase shifts are measured as deviations from the low level case. note that the signal levels over which the input is swept must remain within the dynamic range of the ad8302 for proper operation. c l p gain p a a in nom opt += + ? rev. b
ad8302 20 reflectometer the ad8302 can be configured to measure the magnitude ratio and phase difference of signals that are incident on and reflected from a load. the vector reflection coefficient, , is defined as, = = ? () + () re flected voltage / incident voltage z z / z z lo lo (17) where z l is the complex load impedance and z o is the charac- teristic system impedance. the measured reflection coefficient can be used to calculate the level of impedance mismatch or standing wave ratio (swr) of a particular load condition. this proves particularly useful in diag- nosing varying load impedances such as antennas that can degrade performance and even cause physical damage. the vector reflectom eter arrangement given in figure 13 consists of a pair of directional couplers that sample the incident and reflected sig- nals. the attenuators reposition the two signal levels within the dynamic range of the ad8302. in analogy to equations 15 and 16, the attenuation factors and coupling coefficients are given by: clp p b b in opt += ? (18) clp p a a in nom opt += + ? (19) where nom is the nominal reflection coefficient in db and is negative for passive loads. consider the case where the incident signal is 10 dbm and the nominal reflection coefficient is 19 db. as shown in figure 13, using 20 db couplers on both sides and 30 dbm for p opt , the attenuators for channel a and b paths are 1 db and 20 db, respectively. the magnitude and phase of the reflection coefficient are available at the vmag and vphs pins scaled to 30 mv/db and 10 mv/degree. when is 19 db, the vmag output is 900 mv. the measurement accuracy can be compromised if board level details are not addressed. minimize the physical distance between the series connected couplers since the extra path length adds phase error to . keep the paths from the couplers to the ad8302 as well matched as possible since any differences introduce measurement errors. the finite directivity, d, of the couplers sets the minimum detectable reflection coefficient, i.e., | min (db)|<|d(db)|. source 1db c1c4 c6c5 vp c7 r4 r1 r2 20db incident wave reflected wave z load r5 r6 1 comm mflt 14 inpa vmag 21 3 ofsa mset 31 2 vpos vref 41 1 ofsb pset 51 0 inpb vphs 69 comm pflt 78 ad8302 c2 c8 c3 figure 13. using the ad8302 to measure the vector reflection coefficient off an arbitrary load rev. b
ad8302 21 table iii. evaluation board configuration options component function default condition p1 power supply and ground connector: pin 2 vpos and pins 1 and 3 ground. not applicable r1, r2 input termination. provide termination for input sources. r1 = r2 = 52.3 ? (size 0402) r3 vref output load. this load is optional and is meant to allow the user to simulate r3 = 1 k ? (size 0603) their circuit loading of the device. r5, r6, r9 snubbing resistor r5 = r6 = 0 ? (size 0603) r9 = 0 ? (size 0603) c3, c7, r4 supply decoupling c3 = 100 pf (size 0603) c7 = 0.1 f (size 0603) r4 = 0 ? (size 0603) c1, c5 input ac-coupling capacitors c1 = c5 = 1 nf (size 0603) c2, c8 video filtering. c2 and c8 limit the video bandwidth of the gain and phase c2 = c8 = open (size 0603) output respectively. c4, c6 offset feedback. these set the high-pass corner of the offset cancellation loop and thus with the input ac-coupling capacitors the minimum operating frequency. c4 = c6 = 1 nf (size 0603) sw1 gset signal source. when sw1 is in the position shown, the device is in gain sw1 = installed measure mode; when switched, it operates in comparator mode and a signal must be applied to gset. sw2 pset signal source. when sw2 is in the position shown, the device is in phase sw1 = installed measure mode; when switched, it operates in comparator mode and a signal must be applied to pset. figure 15a. component side metal of evaluation board figure 15b. component side silkscreen of evaluation board table ii. p1 pin allocations 1 common 2 vpos 3 common r5 gain sw1 gset vref pset phase r9 r3 r6 sw2 mflt 14 vmag 13 mset 12 vref 11 pset 10 vphs 9 pflt 8 ad8302 c2 c8 r7 r8 c7 vp vp 1 comm inpa 2 ofsa 3 vpos 4 ofsb 5 inpb 6 comm 7 c1 c4 c6 c5 r4 c3 inpa inpb r1 r2 gnd figure 14. evaluation board schematic rev. b
ad8302 22 characterization setups and methods the general hardware configuration used for most of the ad8302 characterization is shown in figure 16. the characterization board is similar to the customer evaluation board. two reference- locked r and s smt03 signal generators are used as the inputs to inpa and inpb, while the gain and phase outputs are monitored using both a tds 744a oscilloscope with 10 high impedance probes and agilent 34401a multimeters. gain the basic technique used to evaluate the static gain (vmag) performance was to set one source to a fixed level and sweep the amplitude of the other source, while measuring the vmag output with the dmm. in practice, the two sources were run at 100 khz frequency offset and average output measured with the dmm to alleviate errors that might be induced by gain/phase modulation due to phase jitter between the two sources. the errors stated are the difference between a best fit line calcu- lated by a linear regression and the actual measured data divided by the slope of the line to give an error in v/db. the referred to 25 c error uses this same method while always using the slope and intercept calculated for that device at 25 c. response measurement made of the vmag output used the configuration shown in figure 17. the variable attenuator, alpha ad260, is driven with a hp8112a pulse generator pro- ducing a change in rf level within 10 ns. noise spectral density measurements were made using a hp3589a with the inputs delivered through a narda 4032c 90 phase splitter. to measure the modulation of vmag due to phase variation again the sources were run at a frequency offset, f os , effectively creating a continuous linear change in phase going through 360 once every 1/f os seconds. the vmag output is then measured with a dso. when perceivable, only at high frequencies and large input magnitude differences, the linearly ramping phase creates a near sinusoid output riding on the expected vmag dc output level. the curves in tpc 24 show the peak-to-peak out- put level measured with averaging. phase the majority of the vphs output data was collected by generating phase change, again by operating the two input sources with a small frequency offset (normally 100 khz) using the same configuration shown in figure 16. although this method gives excellent linear phase change, good for measurement of slope and linearity, it lacks an absolute phase reference point. in the curves showing swept phase, the phase at which the vphs is the same as vphs with no input signal is taken to be 90 and all other angles are references to there. typical performance curves show two figures of merit; instantaneous slope and error. instanta- neous slope, as shown in tpcs 43, 44, 45, and 47, was calculated simply by taking the delta in vphs over angular change for adjacent measurement points. tektronix tds 744a oscilloscope multimeter/ oscilloscope inpa inpb vmag vref vphs evb 3db r & s signal generator smto3 tektronix vx1410a 3db r & s signal generator smto3 hp 34401a multimeter same setup as v mag figure 16. primary characterization setup inpa inpb vmag vref vphs evb 3db r & s signal generator smto3 splitter variable atten fixed atten tektronix vx1410a 3db p tektronix tds 744a oscilloscope pulse generator figure 17. vmag dynamic performance measurement setup rev. b
data sheet ad8302 rev. b | page 23 of 23 outline dimensions compliant to jedec standards mo-153-ab-1 061908-a 8 0 4.50 4.40 4.30 14 8 7 1 6.40 bsc pin 1 5.10 5.00 4.90 0.65 bsc 0.15 0.05 0.30 0.19 1.20 max 1.05 1.00 0.80 0.20 0.09 0.75 0.60 0.45 coplanarity 0.10 seating plane figure 18. 14-lead thin shrink small outline package [tssop] (ru-14) dimensions shown in millimeters ordering guide model 1 temperature range package description package option ad8302aruz ?40c to +85c 14-lead thin shrink small outline package [tssop] ru-14 ad8302aruz-reel ?40c to +85c 14-lead thin shrink small outline package [tssop] ru-14 ad8302aruz-rl7 ?40c to +85c 14-lead thin shrink small outline package [tssop] ru-14 AD8302-EVALZ 1 z = rohs compliant part. revision history 4/2018rev. a to rev. b updated outline dimensions ........................................................ 23 moved ordering guide .................................................................. 23 changes to ordering guide ........................................................... 23 7/2002rev. 0 to rev. a. tpc 3 through tpc 6 replaced .................................................... 6 ?2018 analog devices, inc. all rights reserved. trademarks and registered trademarks are the prop erty of their respective owners. d02492-0-4/18(b)

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